1. Field
Embodiments of the present invention relate to a switching power supply in which the power factor in the region of small phase angle of the input voltage does not decrease even in the state with a maximum frequency limitation on the switching frequency under a light load condition.
2. Description of the Related Art
FIG. 6 shows a schematic construction of an example of a switching power supply having a power factor correction function. The switching power supply basically comprises an inductor L connected at a terminal thereof to a diode bridge circuit DB that is a rectifier circuit performing full-wave rectification of input AC voltage AC and a switching element Q connected to the other terminal of the inductor L and forming a current path from the diode bridge circuit DB through the inductor L on turning ON of the switching element Q. The switching element Q is a MOS-FET, for example.
The switching power supply further comprises a diode D connected to the other terminal of the inductor L and forming a current path from the inductor L to an output capacitor Cout on turning OFF of the switching element Q to deliver a specified output DC voltage Vout on the output capacitor Cout. The output DC voltage Vout is supplied through an output terminal OUT to an electronic device (not shown in the figure). The above described construction having a main component of the switching element Q composes the main body of a switching power supply of a booster type that obtains an output DC voltage Vout of about 400 V, for example, from AC 100 V.
The control circuit CONT, which is an integrated circuit performing various functions with a monolithic structure, conducts ON/OFF driving the switching element Q to control the current that flows through the inductor L. Specifically, the control circuit CONT receives at a terminal FB thereof a voltage Vfb that is detected through a voltage dividing resistors R4 and R5 and proportional to the output DC voltage Vout. An error detector 11 provided in the control circuit CONT generates a difference voltage between this voltage Vfb and a predetermined reference voltage Vref. The error detector 11 is a transconductance amplifier, for example. The control circuit CONT comprises a comparator 12, which is an overvoltage detector that compares the voltage Vfb received through the terminal FB with a predetermined reference voltage Vovp to detect an overvoltage.
The control circuit CONT receives at a terminal IS thereof a voltage Vis that is detected with a resistor R3 series-connected to the source terminal of the switching element Q and proportional to the current running through the switching element Q. A comparator 13, which is an overcurrent detector (or an overcurrent detecting means) provided in the control circuit CONT, compares this voltage Vis with a predetermined reference voltage Vovc to detect an overcurrent. The control circuit receives at a terminal ZCD thereof through a resistor R2 a winding voltage that is generated across an auxiliary winding La of the inductor L and corresponds to the current through the inductor L. A comparator 14, which is a zero current detector provided in the control circuit CONT, compares this winding voltage with a predetermined reference voltage Vzcd to detect a zero current.
A comparator 15, which is a PWM device or an ON width control means, compares a saw-tooth wave generated by an oscillator 16 with the error output of the error detector 11, the error output being the voltage difference between the voltage Vfb and the reference voltage Vref. This comparator 15 reverses the output thereof into an H level when the saw-tooth wave reaches the error output of the error detector 11, and thus resets a flip-flop 18 through an OR circuit 17. The flip-flop 18 is set by the output of the comparator 14 through an OR circuit 19 when the comparator 14 detects a zero current. The output through the OR circuit 19 also triggers the generation of the saw-tooth wave in the oscillator 16.
The switching element Q is ON/OFF-controlled by the output of the flip-flop 18 that is set and reset as described above. Specifically, the flip-flop 18 controls the gate voltage of the switching element Q giving a set output to a driver circuit not shown in the figure. Thus, the switching element Q is switching-driven in such a way that the switching element Q is turned ON upon setting of the flip-flop 18 and turned OFF upon resetting of the flip-flop 18.
The comparator 15 resets the flip-flop 18 corresponding to the error output of the error detector 11, thereby controlling an ON width of the switching element Q. The comparator 14 sets the flip-flop 18 upon detection of a zero current and simultaneously triggers generation of oscillation in the oscillator 16, thereby controlling ON/OFF period or a switching frequency of the switching element Q.
A resistor R1 connected to the terminal RT regulates a slope of the saw-tooth wave generated by the oscillator 16. Capacitors C1 and C2 and a resistor R6 that are connected to a terminal COMP is a phase regulation circuit for the error output of the error detector 11. The flip-flop 18 is forcedly reset through the OR circuit 17 when an overvoltage is detected by the comparator 12 or an overcurrent is detected by the comparator 13. A timer circuit 10 counts a certain time at the start of the power supply and sets the flip-flop 18 through the OR circuit 19.
In the switching power supply with the construction described above, when the magnitude of the load connected to the output terminal OUT is constant, the error output of the error detector 11 is constant and the switching element Q is switching-controlled with a constant ON width. The input voltage Vin is obtained by full-wave rectification of the input AC voltage AC through the diode bridge DB, and the voltage across the inductor L varies with the phase angle as shown by the waveform (a) in FIG. 7.
The waveform (b) in FIG. 7 is a saw-tooth wave generated in the oscillator 16, which is compared with the error output of the error detector 11 in the comparator 15. The comparison result ON/OFF-controls the switching element Q as shown by the waveform (c) in FIG. 7. The waveform (d) in FIG. 7 shows the current flowing through the inductor L in the ON/OFF operation of the switching element Q. It is apparent that the slope of the inductor current varies depending on the phase angle of the input voltage Vin. The envelope of the peak values of the inductor current forms an AC waveform similar to the input voltage Vin, the peak values being the inductor current at the instances of turning OFF of the switching element Q.
This causes variation in the period of time from the moment of turning OFF of the switching element Q to the moment of zero current through the inductor L. If the switching element Q would be ON/OFF-controlled with a constant period or constant frequency despite this variation, the switching element Q is turned ON under the condition of subjecting to a certain voltage on the switching element Q. Thus, the switching element Q suffers from a significant switching loss.
The zero current detection mentioned earlier detects the moment of zero current through the inductor L after turning OFF of the switching element Q. Zero current switching conducts turning ON of the switching element Q at this moment of zero current, thereby reducing the switching loss in the switching element Q caused by the variation of the peak values of the inductor current.
Patent Documents 1 and 2 disclose in detail basic construction and switching control of this type of switching power supply.    [Patent Document 1]    U.S. Pat. No. 6,984,963    [Patent Document 2]    U.S. Pat. No. 7,116,090
In the switching control described above, the switching frequency Fc increases as the load becomes lighter as shown in FIG. 8. If the switching frequency Fc exceeds the operation frequency of the switching element Q, the switching loss increases. Accordingly, control is conducted to limit the maximum frequency Fmax of the switching frequency Fc as disclosed in Patent Document 2, for example. The limiting the switching frequency Fc below the maximum frequency Fmax is actually carried out by delaying the turning ON timing of the switching element Q, thereby avoiding power factor degradation.
However, there is the following problem in the power factor correction by limiting the maximum frequency of the switching frequency Fc. In the ON/OFF control of the switching element Q as described above, the magnitude of the inductor current is controlled corresponding to the phase angle of the input voltage Vin. Consequently, the inductor current is small in the region of small phase angles. When the maximum frequency limitation of the switching frequency is added under a light load condition in particular, the inductor current hardly flows in the region of small phase angle of the input voltage Vin, resulting in distortion in the input current waveform, which causes degradation of power factor in the region of small phase angles.